Due to that the power density of the power supply and the efficiency of the circuit are continuously increased, the resonant converters are becoming more and more valued for their relatively higher efficiencies and the synchronous rectification technology are commonly applied due to its relatively lower turn-on loss.
But there are certain problems regarding using the synchronous rectification technology in the resonant converters. These problems are addressed by using FIGS. 1 to 5 as examples. As shown in FIG. 1, it is a circuit diagram of a DC/DC full-bridge LLC resonant converter 10 in the prior art. In which, it receives an input voltage Vin for generating an output voltage Vout, and includes an input capacitor C1, a full-bridge switching circuit having four switching switches (Q1-Q4), a resonant tank having a resonant inductor Lr, a resonant capacitor Cr and a magnetizing inductor Lm, wherein these three elements are coupled to each other in series, a transformer T having a primary winding and a secondary winding with a central tap, a full-wave rectifier having two rectifying switches (Q5-Q6) and an output capacitor C2. FIG. 2 shows the main operating waveforms of the DC/DC full-bridge LLC resonant converter as shown in FIG. 1. In which, VQ1, VQ4, VQ2 and VQ3 are gate driving signals of the four switching switches Q1, Q4, Q2 and Q3 respectively; iLm is the waveform of the magnetizing current; iLr is the waveform of the resonant current; iQ5 and iQ6 are the current waveforms of the rectifying switches Q5 and Q6 of the secondary side of the transformer, which are approximate to the sine waveform when Q5 and Q6 are turned on; VQ5 and VQ6 are gate driving signals of the ideal rectifying switches Q5 and Q6, wherein the falling edges of the two gate driving signals VQ5 and VQ6 are guaranteed that Q5 and Q6 are turned off at the zero current crossing timing point that is—the timing point that their currents are crossing zero point towards the negative direction. Viewing from FIG. 2, there are no direct corresponding relationships among VQ5/VQ6, and gate driving signals VQ1, VQ4, VQ2 and VQ3, thus VQ5/VQ6 can not be simply composed by using the primary side driving signals VQ1, VQ4, VQ2 and VQ3, which results in the complexity of the synchronous rectification of the resonant converter.
There are plenty of controlling methods for the resonant converters, one of which is the current sensing controlling method. In FIG. 3, it is a schematic circuit diagram of a DC/DC full-bridge LLC resonant converter system having a synchronous rectification controller in the prior art, which utilizes a zero current crossing comparing and controlling method to obtain the synchronous rectification control. Except for the DC/DC full-bridge LLC resonant converter 10, it further includes a first and a second synchronous rectification controller 11 and 12 and a main controller 13 at the primary side of the transformer T. Since the controlling principles and the structures of the first and the second synchronous rectification controllers 11 and 12 are all the same, the controlling principle and the structure of the second synchronous rectification controller 12 is described as an example. In this controlling method, a turn-on current signal VC6 of the rectifying switch Q6 representing the turn-on current of Q6 is generated via a current transformer (CT) 120 and a current sensor 121 firstly. The turn-on current signal VC6 compares with a reference voltage Vref (Vref=0) via a comparator 122 to obtain a zero crossing signal VCm6 wherein the timing of the falling edge of VCm6 is the same as zero current crossing point that is the timing point when VC6 equals to 0. Then the zero crossing signal VCm6 goes through a processor 123 and a driver 124 to generate the driving signal VQ6 to drive the switch Q6, wherein when VCm6 goes through the processor 123, it engages a series of processes regarding preventing from jitter and reshaping etc. And the first synchronous rectification controller 11 and the second synchronous rectification controller 12 have the same elements, i.e. a CT, a current sensor, a comparator, a processor and a driver, wherein a turn-on current signal of the rectifying switch Q5 is VC5, and a zero crossing signal VCm5 is obtained by sending VC5 through its comparator (which are not shown). The driver of the first synchronous rectification controller 11 generates the driving signal VQ5 to drive the switch Q5. And the main controller 13 generates a first driving signal (which includes gate driving signals VQ1 and VQ4) to drive the first and the fourth switches Q1 and Q4 and a second driving signal (which includes gate driving signals VQ2 and VQ3) to drive the second and the third switches Q2 and Q3. As shown in FIG. 3, the method for sensing the current is to sense the current of the rectifying switch directly through the CT, and there are many other current sampling methods in actual applications, e.g., sensing the switch current via a resistor connected in series with the switch, directly sensing the voltage across two terminals of the switch or sensing a signal in the circuit reflecting the switch current e.g. the primary side current of the transformer which omitting a portion related to the magnetizing current.
This kind of zero crossing comparing and controlling method has simple principles and is easy to accomplish. But there is propagation delay in the circuit so that when the turn-on current equals to zero and even less than zero, VQ6 is still on high level which means Q6 is still on, thus a current reverse-flow phenomenon occurs and a relatively large voltage spike across the corresponding rectifying switch generates. One solution is to increase the comparison threshold value Vref of the comparator 122 such that the timing of the falling edge of the turn-off signal VCm6 is a little bit earlier than the zero current crossing point to counteract the propagation delay. Similarly, the same problem exists in the control method for the rectifying switch Q5 driven by the first synchronous rectification controller 11, and can be solved by the same method. FIG. 4 shows corresponding waveforms of the resonant converter system in FIG. 3. Comparing the non-zero comparison threshold value Vref with the synchronous rectifying switches' turn-on current signals VC5 and VC6 to obtain turn-off signals VCm5 and VCm6 whose falling edge are a little bit earlier than the zero current crossing point that the current signals VC5 and VC6 equals to zero from the positive direction such that the rectifying switches Q5 and Q6 are turned off when the current is at the zero as shown by the waveforms of VQ5 and VQ6 in FIG. 4.
But, this method also has its problems. For different loads, the delay time between the falling edge of the turn-off signal and zero current crossing point are not the same due to the variations of the rectifying currents' waveforms. As shown in FIG. 5, the sampling current signals VC5a (VC6a) and VC5b (VC6b) are different when the loads are different, which results in the falling edges of the driving signals VQ5a and VQ6a generated by comparing the large current signals VC5a and VC6a with the comparison threshold value Vref lagged than the falling edges of the driving signals VQ5b and VQ6b generated by comparing the small current signals VC5b and VC6b with the comparison threshold value Vref. Thus, if the threshold value suitable for the heavy load circumstances is employed at light load condition, the falling edges of the driving signals VQ5b and VQ6b are early than the zero crossing point and part of the current flows through the parasite diodes of the rectifying switches Q5 and Q6, thus the efficiency is decreased. Similarly, if the threshold value suitable for the light load circumstances is employed at heavy load condition, the falling edges of the driving signals VQ5a and VQ6a are latter than the zero crossing point and the current flows reversely, thus the voltage spike of the rectifying switches Q5 and Q6 are generated.
Thus, the best method is to employ an adaptive comparison threshold value for different load current to make sure that the falling edge of the driving signal of the synchronous rectifying switch is at the same timing point e.g. the zero current crossing point at different load condition. The present invention is proposed according to such an idea.
Keeping the drawbacks of the prior arts in mind, and employing experiments and research full-heartily and persistently, the applicant finally conceived a resonant converter system having a synchronous rectification control circuit and a controlling method thereof.